Voltage regulation is commonly required to prevent variation in the supply voltage powering various microelectronic components such as digital ICs, semiconductor memories, display modules, hard disk drives, RF circuitry, microprocessors, digital signal processors and analog ICs, especially in battery-powered applications such as cell phones, notebook computers and consumer products.
Since the battery or DC input voltage of a product often must be stepped-up to a higher DC voltage, or stepped-down to a lower DC voltage, such converters are referred to as DC-to-DC converters. Step-down converters, commonly referred to as “Buck converters,” are used whenever a battery's voltage is greater than the desired load voltage. Step-down converters may comprise inductive switching converters, capacitive charge pumps, and linear converters. Conversely, step-up converters, commonly referred to as “boost converters,” are needed whenever a battery's voltage is lower than the voltage needed to power its load. Step-up converters may comprise inductive switching converters or capacitive charge pumps.
Another type of converter may operate as either a step-up or a step-down converter depending on whether the power input to the converter has a voltage above or below its output voltage. Commonly referred to Buck-boost converters, such circuitry is needed whenever a converter's input and output are similar in voltage, where variations in the input voltage preclude the use of a simple boost or Buck converter.
One example of such an application requiring both step-up and step-down conversion is supplying a regulated 3.3V output from a lithium ion (Lilon) battery. A Lilon battery exhibits a terminal voltage which decays from 4.2V when fully charged to below 3V when discharged. Since the initial battery voltage is above 3.3V and the final battery voltage is below 3.3V, the converter must be able to step-down initially and step-up later.
Inductive Switching Converters
Of the aforementioned voltage converters, the inductive switching converter can achieve superior performance over the widest range of currents, input voltages and output voltages. The operating principles of the inductive switching converter are described in application Ser. No. 11/890,818, titled “High-Efficiency DC/DC Voltage Converter Including Down Inductive Switching Pre-Regulator And Capacitive Switching Post-Converter,” filed contemporaneously herewith and incorporated herein by reference. Two examples of non-isolated inductive switching converters, a synchronous Buck step-down converter and synchronous boost step-up converter, are shown in FIGS. 1A and 1B.
An illustrative synchronous Buck converter 1, shown in FIG. 1A, comprises a power MOSFET switch 3, an inductor 5, a synchronous rectifier power MOSFET 4 with rectifier diode 8, and a capacitor 6. Operation of MOSFET 3 is controlled by a pulse-width modulation (PWM) control circuit 2, driving the gate of MOSFET 3. The gate drive may vary in polarity and voltage depending on whether MOSFET 3 is N-channel or P-channel. Synchronous rectifier MOSFET 4, generally an N-channel MOSFET, is driven out of phase with MOSFET 3, but MOSFET 4 is not necessarily on the entire time when MOSFET 3 is off. In general, MOSFET 4 conducts only during times when diode 8 is conducting.
While the control circuit controlling the converter's operation is referred to as a “PWM control,” implying fixed-frequency variable-pulse-width operation, it may alternatively operate in a variable frequency mode where the clock period is allowed to vary, or alternatively it may alternate between varying and fixed frequency modes depending on load and input conditions.
The energy delivered from the power source, battery or power input into the DC/DC converter 1 is switched or gated through MOSFET 3. With its positive terminal connected to the battery or input, MOSFET 3 acts like a “high-side” switch controlling the current in inductor 5. Diode 7 is a P-N junction parasitic to MOSFET 3, in parallel to the transistor's drain and source, which remains reverse biased under normal Buck converter operation. Since diode 7 does not carry current under normal operation, it is illustrated by dotted lines.
By controlling the current in the inductor 5 through the switching and on-time control of MOSFET 3, the energy stored in the inductor coil's magnetizing field can be adjusted dynamically to control the voltage on output filter capacitor 6. The output voltage Vout is fed back to the input of PWM control circuit 2, which controls the current IL in inductor 5 through the repeated switching of MOSFET 3. The electrical load connected to the converter's output is not shown.
Driven out of phase with MOSFET 3, synchronous rectifier MOSFET 4 conducts some portion of the time when MOSFET 3 is off. With its positive terminal connected to the inductor 5, i.e. to node Vx, and its negative terminal connected the circuit ground, MOSFET 4 acts like a “low-side” switch, shunting the current flowing through diode 8. Diode 8 is a P-N junction parasitic to synchronous rectifier MOSFET 4, in parallel to the transistor's drain and source. Diode 8 conducts substantial inductor current only during intervals when both MOSFETs 3 and 4 are turned off.
One common condition when both MOSFETs 3 and 4 are off occurs during every switching transition to prevent shorting of the input power source to circuit ground. This so-called break-before-make interval prevents shoot through conduction by guaranteeing that both transistors do not conduct simultaneously and short or “crow-bar” the converter's input and power source to ground.
During this brief break-before-make (BBM) interval, diode 8 in parallel to synchronous rectifier MOSFET 4 must, along with any parasitic capacitance associated with diode 8, carry the inductor's current IL. Unwanted noise can occur during the transitions associated BBM operation.
If we define the converter's duty factor D as the time during which energy flows from the battery or power source into the DC/DC converter, i.e. the time that MOSFET switch 3 is on, then the output to input voltage ratio of the Buck converter 1 is proportionate to its duty factor, i.e.
            V      out              V      in        =      D    ≡                  t        sw            T      where tsw is the time during which MOSFET 3 is turned on and T is the duration of the total clock cycle.
This relationship for a Buck or synchronous Buck converter is illustrated by curve 17 in graph 15 of FIG. 2A. Notice the Buck converter cannot smoothly reach a zero or unity transfer characteristic without exhibiting some discontinuities 19 and 21 at the extremes of the duty factor D. This phenomenon is due to switching delays in the power MOSFET switch and its control and gate drive circuitry.
As long as the Buck converter's power MOSFET is still switching, tsw is limited to some portion of the clock period T, e.g. 5%<D<95%, essentially due to turn-on and turn-off delay within the MOSFETs 3 and 4 and their control loop. For example at 95% duty factor and a 3 MHz clock, the off time for the high-side MOSFET 3 is only five percent of the 333 nsec period, or just 16 nsec. This means the high side MOSFET 3 must turn off and back in only 16 nsec—too rapidly to regulate over a 95% output-to-input conversion ratio. The minimum off time problem impacts either synchronous or non-synchronous Buck converters. The problem is, however, further exacerbated in synchronous DC/DC converter 1, since no time remains for the synchronous rectifier MOSFET 4 to turn on and then off again and still exhibit BBM operation.
Referring again to curve 17 in FIG. 2A, above some maximum duty factor Dmax, there is not adequate time to maintain switching operation and the converter must jump from Dmax to a 100% duty factor, as shown by discontinuity 21. Above Dmax, the converter turns MOSFET 3 on and leaves it on for the entire clock period T. The abrupt transition 21 causes a glitch in the output voltage of Buck converter 1. Moreover, at a 100% duty factor, Vout=Vin as shown by line 16, and all regulation is lost as long as the switching is halted.
Synchronous boost converter 10 shown in FIG. 1B includes a low-side power MOSFET 12, a battery connected inductor 13, an output capacitor 15, and a “floating” synchronous rectifier MOSFET 14 with parallel rectifier diode 16. The gates of the MOSFETs 12 and 14 are driven by break-before-make circuitry (not shown) and controlled by PWM controller 11 in response to voltage feedback VFB from the output of converter 10, present across output capacitor 15. BBM operation is needed to prevent shorting the terminals of output capacitor 15.
The synchronous rectifier MOSFET 14, which may be an N-channel or P-channel MOSFET, is considered floating in the sense that its source and drain terminals are not permanently connected to any supply rail, i.e. to ground or Vbatt. Diode 16 is a P-N diode intrinsic to synchronous rectifier MOSFET 14, regardless whether synchronous rectifier MOSFET 14 is a P-channel or an N-channel device. A Schottky diode may be included in parallel with MOSFET 16 but with series inductance may not operate fast enough to divert current from forward biasing intrinsic diode 16. Diode 17, which is a P-N junction diode intrinsic to N-channel low-side MOSFET 12, remains reverse biased under normal boost converter operation. Since diode 17 does not conduct under normal boost operation, it is shown with dotted lines.
If we again define the duty factor D of boost converter 10 as the time during which energy flows from the battery or power source into the converter, i.e. during the time that low-side MOSFET 12 is on and inductor 13 is being magnetized, then the output to input voltage ratio of a boost converter is proportionate to the inverse of 1 minus its duty factor, i.e.
            V      out              V      in        =            1              1        -        D              ≡          1              1        =                              t            sw                    /          T                    where tsw is the time during which MOSFET 12 is turned on and T is the duration of the total clock cycle.
This relationship for a boost or synchronous boost converter is illustrated by curve 18 in FIG. 2A. Notice that the boost converter cannot smoothly reach a unity transfer characteristic without exhibiting some discontinuity at the extremes of D. This phenomenon occurs due to switching delays in the power MOSFET 12 and its control and gate drive circuitry.
As long as power MOSFET 12 is still switching, tsw is limited to some portion of the clock period T, e.g. 5%<D<95%, essentially due to turn-on and turn-off delay within the MOSFET 12 and its control loop. For example at 5% duty factor and a 3 MHz clock, the off time for the low-side MOSFET 12 is only five percent of the 333 nsec period, or just 16 nsec. This means the low side MOSFET 12 must turn on and back off in only 16 nsec—too rapidly to regulate below a 5% output-to-input conversion ratio. This minimum on time problem impacts either synchronous or non-synchronous boost converters.
Referring again to curve 18 in FIG. 2A, below some minimum duty factor Dmin, there is not adequate time to maintain switching operation and the converter 10 must jump from Dmin to a 0% duty factor, as shown by discontinuity 20. Below Dmin, the converter turns on the synchronous rectifier MOSFET 14 and leaves it on for the entire clock period T. The abrupt transition 20 causes a glitch in the output voltage of boost converter 10. Moreover, at a 100% duty factor, Vout=Vin as shown by line 16, all regulation is lost as long as the switching is halted.
So for both synchronous Buck converter 1 and synchronous boost converter 10, operation near a unity transfer characteristic where Vout≈Vin, shown by line 16 in FIG. 2A, is problematic.
The efficiency η of a DC/DC converter can be given by:
  η  =                    P        out                    P        in              =                            I          out                ·                  V          out                                      I          in                ·                  V          in                    
An analysis of inductive switching converter efficiencies is described in the above-referenced application Ser. No. 11/890,818.
Graph 25 of FIG. 2B illustrates examples of typical conversion efficiencies for synchronous Buck and synchronous boost converters as a function of the converter's voltage conversion ratio Vout/Vin. As shown, line 26 illustrates the unity conversion condition where Vout=Vin. Conversion ratios less than unity, on the left side of line 26 in graph 25, represent step-down conversion. Efficiency curve 27 represents an example of a Buck converter performing step-down voltage conversion. Conversion ratios greater than unity, on the right side of line 26 in graph 25 represent step-up conversion. Efficiency curve 28 represents an example of a boost converter performing step-up voltage conversion.
In general boost converters exhibit lower efficiencies than Buck converters for comparable load currents, as illustrated by curves 27 and 28. This disparity is primarily due to the fact that boost converters exhibit higher peak currents than Buck converters. This problem is further accentuated for high Vout/Vin voltage conversion ratios, especially for output voltages approaching ten times their input, as illustrated by the efficiency decline of curve 28 with increasing conversion ratios.
Furthermore, in graph 25, Buck efficiency 27 is not shown for conversion ratios below 0.1 and above 0.9 and likewise boost efficiency 29 is not shown for conversion ratios below 1.1 and above 10, because these ranges require switching converter operation below a 10% or above a 90% duty factor, an operating condition difficult to achieve, especially at high switching frequencies.
Buck-Boost Switching Converter
The problem of non-isolated DC/DC switching converter operation near unity transfer is especially difficult in applications when the input voltage may vary either above or below the desired output voltage. Examples of this application include the output of noisy AC adapters or in circuitry which must operate by battery back-up during emergency conditions when a main source of power has failed.
Another scenario where a unity conversion ratio is required occurs when a battery's operating voltage range extends above and below the desired regulated voltage.
For example, the discharge characteristic of a Lilon battery starts at 4.2V at full charge, initially decays rapidly to around 3.6V, then decays slowly from 3.6V to 3.4V, and finally drops quickly to its cutoff at or below 3V. In the event that a DC/DC converter is needed to produce a well-regulated 3.3V output during the entire duration, a sub-unity conversion ratio of 3.3V/4.2V, or 0.79, is needed at the outset, indicating that a Buck converter is required. At the battery's end-of-life, the conversion ratio exceeds unity becoming 3.3V/3V, or 1.1, requiring a boost converter to achieve regulation. Such an application demanding both step-up and step-down conversion requires a Buck-boost, or up-down converter.
In the case where the user wants to avoid the complexities of up-down conversion, one possible approach is to use only a Buck converter and give up some battery life by cutting of the battery early, e.g. at 3.3V, but in practice when considering battery manufacturing variations and converter drop-out and duty factor limitations, too much battery life is sacrificed to rely on a Buck-only converter solution.
If up-down conversion cannot be avoided, one possible solution involves Buck-boost conversion and regulation. The Buck-boost converter can easily be derived from combining synchronous Buck and boost converters into a merged circuit. In the Buck-boost converter 35 of FIG. 3A, for example, a synchronous Buck converter comprising a P-channel or N-channel MOSFET 36, an inductor 38A, an N-channel synchronous rectifier MOSFET 37, an intrinsic rectifier diode 39, and a capacitor 44 is used to power a synchronous boost converter comprising a low-side N-channel MOSFET 40, an inductor 38B, a synchronous rectifier MOSFET 41, an intrinsic rectifier diode 42, and a filter capacitor 43. Buck-boost converter 35 first steps down the input voltage to an intermediate voltage lower than the desired output, then steps this voltage up to produce Vout.
Conversely, in the synchronous boost-Buck converter 45 of FIG. 3B, a boost converter comprising a low-side N-channel MOSFET 46, an inductor 47, an N-channel or P-channel synchronous rectifier MOSFET 48A, an intrinsic diode 49, and a capacitor 54 is used to power a synchronous Buck converter comprising a MOSFET 48B, an inductor 52, an N-channel synchronous rectifier MOSFET 50, an intrinsic rectifier diode 51, and a filter capacitor 53. The cascade boost-Buck converter 45 drives a load (not shown). In this approach the input voltage is first stepped-up to an intermediate voltage higher than the desired output, then back down to produce Vout.
The overall efficiency of either Buck-boost converter 35 or boost-Buck converter 45 is given by the product of the boost converter's efficiency ηboost multiplied by the Buck converter's efficiency ηBuck, mathematically as ηcascade=ηBuck·ηboost. Even if both converters are 85% efficient, the combined cascade converter only reaches an overall efficiency of roughly 70%, significantly lower than the efficiency of an individual Buck converter or boost converter. The overall loss of either a Buck-boost or boost-Buck cascade is worse than a synchronous Buck or synchronous boost alone, because there are more transistors in series between input and output, and because all the transistors are switching all the time.
As shown, boost-Buck converter 45 of FIG. 3B includes series-connected MOSFETs 48A and 48B with intermediate capacitor 54. Since in steady-state, the current in series-connected MOSFETs must be equal, MOSFET 48B is redundant and can be eliminated without impacting circuit operation. Even so, boost-Buck converter 45 requires two inductors 47 and 52, a characteristic highly undesirable from a user's point-of-view.
Similarly, Buck-boost converter 35 of FIG. 3A includes inductors 38A and 38B with intermediate capacitor 44. Since in steady state the current in inductors 38A and 38B is the same, inductor 38B is redundant and may be eliminated without changing the function of the circuit. In fact, capacitor 44 may also be eliminated without significantly altering the converter's operation.
The resulting simplified prior-art Buck-boost converter 55 is illustrated in FIG. 3C, comprising a single-inductor 59; four MOSFETs 57, 56, 60, and 61; diodes 58 and 62 and filter capacitor 63. The PWM control circuitry and break-before-make and gate buffer circuits are not shown. Depending on its terminal conditions, such a converter can operate in three distinct modes, Buck, boost, and Buck-boost.
In FIG. 3D, schematic diagram 65 represents the operation of Buck-boost converter 55 as a Buck converter, where MOSFETs 57 and 56 are switched out-of-phase by a PWM control unit (not shown), while MOSFET 61 remains turned-on, represented as resistance 67, and MOSFET 60 is turned off, shown as open circuit 66. The overall power loss in converter 55 is greater than the power loss in a synchronous Buck converter because it now includes the conduction loss in MOSFET 61, i.e. power lost continuously in resistance 67. As a result of this increased power loss, Buck-boost converter 55 operating in its Buck mode has a lower efficiency than conventional Buck converter 1 shown in FIG. 1A.
In FIG. 3E, schematic diagram 70 represents the operation of Buck-boost converter 55 as a boost converter, where MOSFETs 60 and 61 are switched out-of-phase under a PWM control unit (not shown), while MOSFET 57 remains turned-on, represented as resistance 71, and MOSFET 56 is turned off, shown as open circuit 72. The overall power loss in converter 55 is greater than the power loss in a synchronous boost converter because it now includes the conduction loss in MOSFET 57, i.e. power lost continuously in resistance 71. As a result of this increased power loss, Buck-boost converter 55 operating in its boost mode has a lower efficiency than conventional boost converter 10 shown in FIG. 1B.
The loss of efficiency using Buck-boost converter 55 is illustrated in FIG. 4 in the plot of efficiency η for various output-to-input voltage conversion ratios Vout/Vin. For convenience, conventional Buck and boost efficiency curves 27 and 28 from FIG. 2B are illustrated by curves 81 and 82 respectively.
Curve 83 illustrates the efficiency of Buck-boost converter 55 operating in Buck-only mode shown in equivalent circuit 65. Because of series resistance 67 associated with on-state MOSFET 61, the efficiency of Buck-boost converter 65 in the Buck only mode is lower than that of a simple Buck converter, represented by curve 81. This loss of efficiency can range from a few percent to over ten percent, depending on operating conditions. Curve 85 illustrates Buck-boost converter 55 operating in full Buck-boost mode where all four switches are switching constantly, and as a result exhibits even greater losses and poorer efficiency than Buck mode curve 83.
Curve 84 illustrates the efficiency of Buck-boost converter 55 operating in boost-only mode shown in equivalent circuit 70. Because of series resistance 71 associated with on-state MOSFET 57, the efficiency of a Buck-boost converter 65 in the boost-only mode is lower than that of a simple boost converter, represented by curve 82. This loss of efficiency can range from a few percent to over ten percent depending on operating conditions. Curve 86 illustrates Buck-boost converter 55 operating in full Buck-boost mode where all four switches are switching constantly, and as a result exhibits even greater losses and poorer efficiency than boost mode curve 84.
Operating near unity conversion ratios, where the output voltage is slightly above or below its input, i.e. where Vout≈Vin, Buck-boost converter 55 must operate in the Buck-boost mode, where all four transistors are switching constantly. The resulting efficiency, represented by curve 87, can be ten to twenty percent lower than the efficiency of conventional Buck and boost converters, represented by curves 81 and 82.
Thus, the efficiency penalty for a prior art Buck-boost converter operating over a wide range of voltage conversion ratios is substantial. Moreover, the converter must change its operating mode whenever operating near unity voltage conversion ratios.
Charge Pump Converters
An alternative to the switched-inductor converter is a charge pump, a voltage conversion circuit using only switches and capacitors to perform voltage translation through repeated charge redistribution, i.e. the continuous charging and discharging of a capacitor network driven by a clock or oscillator.
The advantage of a charge pump is that specific voltage conversion ratios, it can exhibit extremely high conversion efficiencies approaching 100%. The disadvantage is that it can only efficiently generate voltages that are selected multiples of the input voltage, determined by the number of “flying capacitors” used in its circuit. At output voltages other than selected multiples of the input voltage, the charge pump exhibits low efficiencies.
An example of a common charge pump 90 is illustrated in FIG. 5A, where a single “flying capacitor” 93 is employed as a “doubler”, i.e. to double the battery's input voltage. Charge pump 90 comprises four MOSFETs, 92, 91, 94 and 95 configured in an H-bridge arrangement, except that one terminal, the source of MOSFET 95 is connected to the charge pump output VCP and reservoir capacitor 96 rather than to ground.
Operation of charge pump 90 involves repeatedly charging and discharging flying capacitor 93. During the charging phase, diagonal MOSFETs 94 and 91 are turned on, charging capacitor 93 to the voltage Vbatt while MOSFETs 92 and 95 remain turned off. Alternatively, in the charge transfer phase, MOSFETs 94 and 91 are turned off, MOSFETs 92 and 95 are turned on, and energy is transferred from the flying capacitor 93 to the output reservoir capacitor 96, pumping the VCP voltage to a value twice the battery voltage or 2·Vbatt.
The purpose of the switch network is essentially to place the flying capacitor in parallel with the battery during charging and in series, i.e. stacked on top of the battery's positive terminal, during discharging, as illustrated by equivalent circuit 100 in FIG. 5B, where voltage source 101 represents the battery input and capacitor 102 charged to Vbatt represents the flying capacitor. By stacking one voltage atop the other, the output voltage of the charge pump is the sum of the voltages, hence doubling the voltage input. The cycle then repeats with another charging phase.
FIG. 5C illustrates a charge pump 110 utilizing two flying capacitors 114 and 115 and a network of seven MOSFETs 111, 112, 113, 116, 117, 118 and 119. This network charges the flying capacitors 114 and 115 in series, charging each flying capacitor to one-half the battery voltage, i.e. Vbatt/2. During the charging stage, MOSFETs 111, 112 and 113 are turned on and MOSFETs 116, 117, 118 and 119 are turned off. After the charging is completed, the two charged capacitors are connected in parallel to the positive terminal of the battery. This connection is accomplished by turning on MOSFETs 116, 117, 118 and 119. The resulting output, shown in equivalent circuit 121 of FIG. 5D, is then Vbatt+Vbatt/2, for an output voltage of 1.5Vbatt as illustrated by battery voltage source 124 with capacitors 122 and 123 stacked atop one another. Because the output is 1.5 times its input this charge pump is sometimes referred to as a “fractional” charge pump.
Actually many charge pump topologies are possible, but most concentrate on using only one or two flying capacitors. A single flying capacitor charge pump is capable only of efficiently delivering an output voltage equal to twice its input voltage, or alternatively, if the capacitor is connected to the negative terminal of the battery, to produce a mirror-image negative voltage of the battery, i.e. −Vbatt. In this topology, the device is known as an inverter. The inverting case is illustrated in equivalent circuit 130 of FIG. 5E, where battery 131 is used to charge capacitor 132 to a voltage below ground, i.e. a voltage referenced to the negative terminal of battery 131. Two-transistor fractional charge pumps may also be used to produce an output voltage equal to positive or negative one-half the input voltage, as shown in equivalent circuit 135 of FIG. 5F, where capacitors 137 and 138, after being charged to one-half of the battery voltage 136 are then referenced to ground to produce either a positive potential +0.5Vbatt or a negative potential −0.5Vbatt.
The problem with charge pump converters is they operate efficiently only at specific multiples of the number of flying capacitors. In other words, they are not true voltage converters. Specifically, as a desired load voltage Vout drops below the voltage VCp the capacitor network produces, the converter cannot adapt. The voltage-differential between the charge pump's output voltage VCP and the regulated output voltage of the converter Vout requires a resistor or current source to support the voltage mismatch, and the voltage across that lossy element results in lost power and reduced efficiency. An analysis of charge pump efficiencies is described in detail in the above-referenced application Ser. No. 11/890,818.
This efficiency equation for single-mode charge pumps is illustrated graphically in FIG. 6A for various multipliers, including a doubler (curve 151), an inverter (curve 152), and fractional charge pumps (curves 153, 154 and 155). Curve 156 represents a direct battery connection, identical to a linear converter's maximum theoretical efficiency, i.e. assuming no quiescent operating current. In each case, as the input to output ratio approaches an integral multiple of ±½Vbatt, the efficiency increases. Above that voltage, the charge pump is not capable of delivering a higher voltage and a different capacitor multiplier, i.e. a different operating mode must be employed.
Each curve shown in graph 150 of FIG. 6A represents a specific charge pump circuit, e.g. including those shown in FIGS. 5A-5F. Unless a load operates at an exact half-volt integral multiple of the input voltage, however, the efficiency of the charge pump converter using one or two capacitors will suffer. This behavior is especially problematic for battery powered products where the battery voltage changes markedly as the cell discharges. In the case of Lilon batteries, the voltage can decay more than 1V during discharge, representing a 25% change. Even if the peak efficiency may be high at one specific operating condition and battery voltage, the overall efficiency of the converter averaged over the battery discharge curve is poor. Weighted average efficiencies can be lower than 60% using a single-mode charge pump.
One way to improve the average efficiency of the converter is to switch modes between 1×, 1.5× and 2× automatically within one circuit. This feature is particularly useful to supply a fixed voltage over a wide input range. An example of the efficiency of a mode changing charge pump is illustrated in graph 160 of FIG. 6B, which shows the efficiency of a tri-mode converter circuit as it switches from 1×-battery-direct mode having an efficiency shown by line 163, to 1.5×-fractional-mode with efficiency curve 162, and again to 2×-doubler-mode with an efficiency curve 161 as the battery decays. By switching modes in this zigzag pattern, the efficiency of the charge pump converter is improved because the output is not pumped to an excessively high value compared to the load.
Unfortunately, conditions still exist where the efficiency suffers substantially. The mode transitions exhibit dramatic shifts in efficiency shown by curve 163 at a conversion ratio of one, and again by curve 162 at a 1.5× conversation ratio. The mode transitions may also result in sudden current and voltage discontinuities, or produce instability or noise. To determine what conversion ratio is required, graph 160 also includes curves 166, 165, and 164 relating the required input voltage range and conversion ratios to produce an output voltage of 3V, 3.5V and 4V respectively.
Specifically, the charge pump converter in 1.5× mode does not perform well for conditions slightly above a unity conversion ratio, manifesting even lower efficiencies than the aforementioned inductive Buck-boost converter.
Dropout in Prior Art Converters
Whenever the input and the output of a voltage converter approach a range of several hundred millivolts of one another, i.e. Vout≈Vin±200 mV, the quality of the converter's regulating ability suffers. Loss of regulation quality may be manifested in several ways, either by a one-time or repeated glitch or discontinuity in output voltage, by increased ripple, or by complete loss of regulation within some narrow voltage band. The phenomenon of degraded regulation whenever Vout approaches Vin is referred to as “dropout”, meaning the converter drops out of regulation.
As shown in FIG. 2A, the Buck converter 1 of FIG. 1A momentarily loses regulation as its switching duty factor jumps from Dmax to 100% or Dmin to 0% and the boost converter 10 of FIG. 1B momentarily loses regulation as its switching duty fact jumps from Dmin to 0%. Both converters completely lose regulation while D=0% since the input is essentially resistively connected to the output during the dropout condition.
While a Buck-boost converter doesn't really exhibit permanent dropout, it can easily suffer a voltage glitch whenever the converter switches from its Buck mode into its Buck-boost mode, or from its Buck-boost mode to its boost mode. Mode transitions occur whenever the converter changes from a circuit having two power devices switching into one where four devices are switching, or vice versa.
To avoid the mode switching transition problem, a Buck-boost converter can be run continuously in Buck-boost mode with all four power devices switching continuously but, as shown in FIG. 4, when this happens the efficiency is degraded under all input-output conditions and conversion ratios.
As stated above, the charge pump is incapable of regulating voltage without the use of a series connected linear converter to provide the regulation function. Unfortunately, it is well known phenomenon that all linear converters exhibit loss of regulation, i.e. dropout, whenever the ΔV across the linear converter's input and output terminals becomes too small. In essence, dropout occurs in a linear converter because the loop gain of the amplifier performing regulation drops precipitously as its transistor pass element changes from behaving as a current source into acting like a variable resistor. If the pass element is a bipolar transistor, the loss of gain occurs at small values of VCE as the device transitions from its active operating region into saturation. In many bipolar linear converters, this dropout condition occurs at more than 400 mV.
In so-called “low dropout” linear converters or “LDOs”, a MOSFET capable of operating as a current source at a lower ΔV is substituted for the bipolar pass element, but the linear converter still drops out at 200 to 300 mV as the power MOSFET pass element transitions from its saturation, i.e. constant current, region into its linear, i.e. resistive, region of operation.
In conclusion, all prior-art non-isolated high-efficiency converters exhibit dropout at voltage conversion ratios approaching unity. Mode switching, loss of regulation and dropout can be avoided, but only by sacrificing efficiency. Isolated converters such as the flyback and forward converter are able to operate at high efficiencies near unity conversion without the need switching modes, but their use of physically-large tapped inductors, coupled inductors, and transformers precludes their application in most portable products.
Summary of Prior-Art Down-Up Converters
In conclusion, no existing charge pump converter, Buck-boost switching converter or other inductive switching converter is able to both step-up and step-down DC voltages efficiently, especially for conversion ratios near unity, where Vin≈Vout. What is needed is an up-down converter that is efficient over a wide range of input and output voltages, and that does not need to change its operating mode as it approaches or operates near unity voltage conversion ratios. Furthermore, the converter should be free from dropout problems, maintaining high quality regulation even while biased with an output voltage within 200 mV of its input, i.e. within the range Vout ≈±200 mV.